Signal processing circuit for floating signal sources using positive feedback

ABSTRACT

A signal processing circuit using positive feedback while keeping the open loop gain of the circuit less than 1 to avoid oscillation. The circuit includes a floating signal source, a low gain amplifier, a feedback element, and a second stage circuit. The floating signal source produces a voltage that is impressed across the feedback element by the feedback system. The feedback element processes the voltage into an output current. The output current is passed through an output current node to the second stage circuit where the output current can be used as a current reference or be further processed. The output from the low gain amplifier may be used as a voltage output node that provides a voltage that is an amplification of the voltage produced by the floating signal source. The signal processing circuit may be embedded in another circuit, including additional stages of the signal processing circuit.

This is a divisional of application Ser. No. 09/925,176, filed Aug. 8,2001.

The present invention relates generally to the field of signalprocessing and particularly to a signal processing circuit for floatingsignal sources using positive feedback.

BACKGROUND OF THE INVENTION

Floating source signals are normally processed in order to makeeffective use of the source signal. Normally, negative feedback systemsare used to process floating source signals, to linearize the transferfunction, and to stabilize the characteristics of the circuit.

Referring to FIG. 1A, a system diagram for a basic feedback system isshown. The feedback system can be implemented with negative or positivefeedback to process floating source signals. The feedback system iscomprised of a floating signal source 1 coupled to a forward gaincircuit 2 and a feedback gain circuit 3. The forward gain circuit 2 hasa forward gain of A and includes an input terminal 2 a and an outputterminal 2 b. The feedback gain circuit 3 has a feedback gain of F andincludes an input terminal 3 a and an output terminal 3 b. The gain ofthe feedback system is calculated as the ratio of voltage at the outputterminal 2 b to the voltage at the input terminal 2 a of the forwardgain circuit 2.

Referring to FIG. 1B, a circuit diagram for a conventional negativefeedback circuit is shown. A signal from a floating signal source 4 iscoupled to a forward gain circuit 9, which in turn is coupled to afeedback gain circuit 6. The forward gain circuit 9 is comprised of anamplifier 5 with a first input terminal 5 a coupled to the floatingsignal source 4, a second input terminal 5 b coupled to the feedbackgain circuit 6, and an output terminal 5 c coupled to the feedback gaincircuit 6. The feedback gain circuit 6 includes a first resistor R1 7and a second resistor R2 8 that are both coupled to the second inputterminal 5 b of the amplifier 5. The floating signal source 4 produces avoltage V_(in). The forward gain circuit 9 has a forward gain of A andthe feedback gain circuit 6 has a feedback gain of F.

The gain of the conventional negative feedback circuit is calculated asthe ratio of voltage V_(out) at the output terminal 5 c of the amplifier5, to the voltage V_(in) produced by the floating signal source 4 at thefirst input terminal 5 a of the amplifier 5. For negative feedbacksystems, the system gain V_(out)/V_(in) is equal to A/(1+FA). Theequation for the system gain of a negative feedback system is shown bythe following equations and analysis. Referring to FIG. 1B:

F=R 2/(R 1+R 2)  1

where F is the feedback gain of the feedback gain circuit 6, R1 is thevalue of the first resistor 7, and R2 is the value of the secondresistor 8.

G=V _(out) *F  2

where G is the voltage at point G and V_(out) is the voltage at theoutput terminal 5 c of the amplifier 5.

V _(out)=(V _(in) −G)*A  3

where V_(in) is the voltage produced by the floating signal source 4 andA is the the forward gain of the amplifier 5. Therefore, using equations2 and 3:

V _(out) =[V _(in)−(V _(out) *F)]*A

V _(in) *A=V _(out) *(1+FA)

V _(out) /V _(in) =A/(1+FA)

The forward gain A is typically much greater than 1 for negativefeedback systems, and can often be in the range of 10⁵ or 10⁶.Therefore, the system gain can be approximated as A/FA or 1/F.

If, for example, the feedback gain circuit 6 consists of a resistordivider including the first resistor R1 7 with a resistance value of 9Rand the second resistor R2 8 with a resistance value of 1R, then thefeedback gain F would be equal to 1/10, thereby, producing a system gainof 1/F or 10. By choosing appropriate resistance values, other valuesfor the feedback gain F and the system gain may be obtained.

High gain operational amplifiers are commonly used as the forward gaincircuit in negative feedback systems. However, operational amplifiersare complex and slow since a signal must pass through several stages oftransistors. The speed of the circuit is degraded with each stage. Thespeed of operational amplifiers is also degraded by the integratingcapacitor used to stabilize the amplifier.

Another disadvantage of negative feedback systems is that they are proneto instability or oscillation. Oscillation will occur if there is afrequency where the open loop gain of the feedback system is greaterthan one and the signal passing through the feedback system is phaseshifted 360 degrees. The open loop gain of the feedback system isdetermined by the product of the forward gain A and the feedback gain F.In negative feedback systems, there is a chance of oscillation since theforward gain A is almost always much greater than 1, and therefore theopen loop gain will almost always be greater than 1.

Also, since the signal inversion of the negative feedback system givesthe equivalent of a 180 degree shift and the operational amplifier usedin negative feedback systems provides a 90 degree phase shift at allfrequencies, if the feedback gain circuit 3 provides an additional 90degree phase shift, the negative feedback system will oscillate.Although a feedback gain circuit consisting of only resistors will notcause a phase shift, any feedback gain circuit containing a capacitormay cause a 90 degree phase shift in the signal and cause oscillation.

Negative feedback circuits can also contribute a significant amount ofnoise due to the resistors typically used in negative feedback circuits.Transistors used in amplifiers also contribute noise. A major source oftransistor noise can be modeled as a current source between thecollector and emitter wherein a larger current contributes more noise.

A final disadvantage of negative feedback systems is that manyoperational amplifiers have a constant gain-bandwidth product (i.e., theproduct of the closed loop gain and the closed loop bandwidth equals aconstant) and therefore must trade-off a high gain characteristic with anarrower bandwidth.

Positive feedback systems, on the other hand, are usually avoided insignal processing because they will oscillate or latch up if the openloop gain is greater than 1, which makes it difficult to design suchsystems to produce an adequate gain. This is because positive feedbacksystems inherently have a 360 degree phase shift.

Therefore, it is an object of this invention to provide a processingcircuit for floating source signals that has greater speed, simplicity,stability, reduced noise, and increased bandwidth over the existingmethods.

SUMMARY OF THE INVENTION

The present invention comprises a positive feedback signal processingcircuit having an open loop gain of less than 1 to avoid oscillation.The signal processing circuit includes a floating signal source, aforward gain circuit comprising a low gain amplifier, and a feedbackgain circuit comprising a feedback element and a second stage circuit.The floating signal source produces a voltage that is impressed acrossthe feedback element by the feedback system. The feedback elementconverts the voltage into an output current. The output current isforced through an output current node to the second stage circuit wherethe output current can be used as a current reference or be furtherprocessed (e.g., amplified to a useable level).

The output from the low gain amplifier may be used as a voltage outputnode that provides a voltage that is an amplification of the voltageproduced by the floating signal source. The signal processing circuitmay be embedded in other circuits, including additional stages of thesignal processing circuit, to create more complex functions and tocreate another stage of gain.

The signal processing circuit of the present invention has several usesincluding use as a floating voltage source amplifier, a currentamplifier, a voltage source to current source converter, and can also beused for analog signal processing functions (e.g., integration,differentiation, exponentiation, and logarithms). The circuits of thepresent invention can be manufactured using gallium arsenide (GaAs)metal-semiconductor field-effect transistor (MESFET) processes andbipolar and complementary metal-oxide silicon or BiCMOS processes.

The signal processing circuit of the present invention is built around ahigh speed, low gain amplifier, providing high speed operation, highbandwidth and low noise. The signal processing circuit of the presentinvention is also completely stable since it operates with an open loopgain of less than 1, thereby avoiding the possibility of oscillation.

BRIEF DESCRIPTION OF THE DRAWINGS

Additional objects and features of the invention will be more readilyapparent from the following detailed description and appended claimswhen taken in conjunction with the drawings, in which:

FIG. 1A is a diagram of a feedback system that can be implemented ineither positive or negative feedback mode.

FIG. 1B is the general circuit layout for a conventional negativefeedback circuit used for signal processing.

FIG. 2 is the general circuit layout for a signal processing circuitusing positive feedback in accordance with the present invention.

FIG. 3 is a diagram of a photo diode amplifier utilizing a photo diodeas the floating signal source and a capacitor as the feedback element inaccordance with the present invention

FIG. 4 is a diagram of a high impedance circuit coupled to the photodiode amplifier of FIG. 3.

FIG. 5 is a diagram of a voltage source to current source converter andvoltage amplifier utilizing a floating voltage source and a resistor asthe feedback element in accordance with the present invention.

FIG. 6 is a diagram of a power supply independent current sourceutilizing a diode set as the floating signal source and a resistor asthe feedback element in accordance with the present invention.

FIG. 7 is a diagram of an embedded circuit utilizing the voltage sourceamplifier of FIG. 5.

FIG. 8 is a diagram of an alternative embodiment of the floating signalsource for use in a current source to current source converter.

FIG. 9 is a diagram of an alternative embodiment of the floating signalsource for use in an integrator circuit.

FIG. 10 is a diagram of an alternative embodiment of the floating signalsource for use in a differentiator circuit.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1A shows a feedback system that can be implemented with negative orpositive feedback to process floating source signals. As stated above,the feedback system is comprised of a forward gain circuit 2 with aforward gain of A coupled to a feedback gain circuit 3 with a feedbackgain of F. The forward gain circuit 2 includes an input terminal 2 a andan output terminal 2 b.

The gain of the feedback system is calculated as the ratio of voltage atthe output terminal 2 b to the voltage at the input terminal 2 a of theforward gain circuit 2. For negative feedback systems, the system gainis equal to A/(1+FA). Negative feedback systems have been extensivelyused in the prior art as described above.

The present invention offers a circuit architecture for implementing apositive feedback system. Referring to FIG. 2, the general circuitlayout for a signal processing circuit using positive feedback inaccordance with the present invention is shown. The signal processingcircuit is built around a forward gain circuit 21 with a forward gain ofA. The forward gain circuit 21 comprises a low gain amplifier 20 with ahigh impedance input terminal 20 a and an output terminal 20 b. Theinput terminal 20 a of the low gain amplifier 20 is coupled to afloating signal source 22 while the output terminal 20 b of the low gainamplifier 20 is coupled to a feedback gain circuit 25 with a feedbackgain of F. The floating signal source 22 produces a voltage V_(in).

The feedback gain circuit 25 is comprised of a feedback element 24 and asecond stage circuit 28. The floating signal source 22 and feedbackelement 24 are coupled at an output current node 26 from which an outputcurrent of I_(out) 26 a flows to the second stage circuit 28.Alternately, or in addition, the output terminal 20 b of the low gainamplifier 20 is coupled to an output voltage node 27 that has anassociated output voltage V_(out).

The system gain (V_(out)/V_(in)) of a basic positive feedback system isequal to A/(1−FA). The equation for the system gain of a positivefeedback system is shown by the following equations and analysis.Referring back to FIG. 1A:

Y=V _(out) *F  1

where Y is the voltage at point Y and V_(out) is the voltage at theoutput terminal 2 b of the forward gain circuit 2.

X=(V _(out) *F)+V _(in)  3

where X is the voltage at point X and V_(in) is the voltage produced bythe floating signal source 1.

V _(out) =X*A  4

where A is the forward gain of the forward gain circuit 2. Therefore,using equations 3 and 4:

V _(out)=[(V _(out) *F)+V _(in) ]*A

V _(in) *A=V _(out)*(1−FA)

V _(out) /V _(in) =A/(1−FA)

As shown in FIG. 1A, the forward gain circuit 2 of a basic negative orpositive feedback system receives two inputs signals: one input signalis received from the floating signal source 1 and another input isreceived from the feedback gain circuit 3.

As shown in FIG. 1B, the forward gain circuit 9 of the conventionalnegative feedback circuit also receives two inputs: one input signal isreceived from the floating signal source 4 and another input signal isreceived from the feedback gain circuit 6. As shown in FIG. 2, theforward gain circuit 21 of the positive feedback circuit of the presentinvention receives only one input signal. Nevertheless, the positivefeedback circuit of the present invention can be made to conform to thebasic positive feedback system model, as shown by the followingequations and analysis. Referring to FIG. 2:

F=R 2/(R 1+R 2)  1

where F is the feedback gain of the feedback gain circuit 25, R1 is theresistance value of the feedback element 24, and R2 is the resistancevalue of the second stage circuit 28.

Y=V _(out) *F  2

where Y is the voltage at point Y and V_(out) is the voltage at theoutput terminal 20 b of the low gain amplifier 20.

X=(V _(out) *F)+V _(in)  3

where X is the voltage at point X and V_(in) is the voltage produced bythe floating signal source 22.

V _(out) =X*A  4

where A is the forward gain of the low gain amplifier 20. Therefore,using equations 3 and 4:

V _(out)=[(V _(out) *F)+V _(in) ]*A

V _(in) *A=V _(out)*(1−FA)

V _(out) /V _(in) =A/(1−FA)

Therefore, as for the basic positive feedback system, the system gain(V_(out)/V_(in)) of the positive feedback circuit of the presentinvention is also equal to A/(1−FA).

If, for example, the feedback gain circuit 25 includes the feedbackelement 24 with a resistance value of 1R and the second stage circuit 28with a resistance value of 9R, the feedback gain F is equal to 9/10. Ifthe forward gain circuit 21 consists of a unity gain amplifier, theforward gain A is equal to 1. These values for the feedback gain F andthe forward gain A correspond to a system gain of 10.

The open loop gain of a feedback system is defined to be the product ofthe forward gain A and the feedback gain F. In the present invention,the forward gain circuit preferably consists of a low gain amplifierwith a gain greater than or equal to 0.5 and less than or equal to 2.Since the forward gain circuit consists of a low gain amplifier, theopen loop gain can easily be kept lower than 1 by using an appropriatefeedback gain circuit. For example, if a low gain amplifier with a gainof 2 is used for the forward gain circuit, the open loop gain can stillbe kept less than 1 by using a feedback gain circuit having a gain ofless than 0.5.

A unity gain amplifier is especially effective in the forward gaincircuit 21 of the present invention since an open loop gain of less than1 can be achieved by combining the unity gain amplifier with anyfeedback gain circuit having a gain of less than 1. Feedback gaincircuits with a gain of less than 1 are easily constructed sincefeedback gain circuits using only passive elements very often have again of less than 1. If the open loop gain of the system is less than 1at all frequencies, then it is impossible for the feedback system (andthus the signal processing circuit) to latch up or oscillate.

The low gain amplifier 20 of the present invention may have a gaingreater than or equal to 0.5 and less than or equal to 2 and may consistof a single transistor (such as an emitter follower), more than onetransistor (such as a Darlington pair), or an operational amplifierconnected as an unity gain buffer. The floating signal source 22 may bea floating voltage source, such as a sensor, microphone, or disk drivesensor. Alternately, the floating signal source may be a floatingcurrent source, such as a photo diode, that is configured (e.g., bycombining it with a capacitor or other circuit element) to produce avoltage. The feedback element 24 may consist of capacitors, inductors,resistors, diodes, or any combination of the above.

If the low gain amplifier 20 is an emitter follower, it will have anoffset voltage between its output and its input. In most embodiments,this offset voltage should be compensated for by using a transistor inseries with the input of the low gain amplifier 20, where the additionaltransistor is configured to compensate for the amplifier's offsetvoltage. A compensating transistor should also be used when a precisioncircuit is desired. In GaAs processes, a typical low gain amplifier hasa voltage offset of 0.3V whereas, in bipolar processes, a typical lowgain amplifier has a voltage offset of 0.6V to 1V. The offset voltagemay be compensated using a transistor to produce an equal offset voltagein the opposite direction of the voltage offset of the low gainamplifier 20. If the feedback element 24 is a capacitor, no compensationis needed since the voltage offset of the low gain amplifier 20 iscanceled by the differentiation function of the capacitor.

In accordance with the present invention, the floating signal source 22produces a voltage V that is impressed across the feedback element 24 bythe feedback system. The feedback element 24 processes the voltage Vinto an output current I_(out) 26 a. The output current I_(out) 26 a ispassed through the output current node 26 to a second stage circuit 28.The type of processing performed on voltage V to produce the outputcurrent I_(out) 26 a depends on the feedback element used.

If the feedback element 24 is a capacitor, then voltage V isdifferentiated to develop the output current I_(out) 26 a, as shownbelow in an embodiment of the present invention. If the feedback element24 is an inductor with an inductance value of L, voltage V is integratedby the feedback element 24 with the value of the output current I_(out)26 a being determined by the equation$I_{out} = {\frac{1}{L}{\int{V.}}}$

If the feedback element 24 is a resistor with a resistance value of R,voltage V is divided by the feedback element 24 with the value for theoutput current I_(out) 26 a being determined by the equationI_(out)=V/R. If the feedback element 24 is a diode, voltage V isexponentiated by the feedback element 24 with the value for the outputcurrent I_(out) 26 a being determined by the equation

I _(out) =I _(S) e ^((qV/kT))

where I_(s) is a constant, q is the charge of an electron, k isBoltzmann's constant, and T is temperature in kelvin.

The output current I_(out) 26 a created by the feedback element 24 doesnot flow back into the floating signal source 22 due to the high inputimpedance of the low gain amplifier 20. Therefore, the output currentI_(out) 26 a flows to a second stage circuit 28, preferably with a lowimpedance, where it can be further amplified to a useable level or beused as a current reference or a bias line since the output currentI_(out) 26 a is a voltage independent current source.

Referring to FIG. 3, a photo diode amplifier 300 in accordance with thepresent invention is shown. The photo diode amplifier has a low gainamplifier 30, which includes an input terminal 30 a and an outputterminal 30 b, a photo diode 32 as a floating signal source, and acapacitor C_(f) 34 as a feedback element. The photo diode 32 isrepresented in FIG. 3 by its equivalent circuit model, which includes acapacitor C_(pd) 32 a in parallel with a current source I_(pd) 32 b. Thephoto diode 32 and capacitor C_(f) 34 are coupled at an output currentnode 36. Although not shown in FIG. 3, an inductor may be coupledbetween the capacitor C_(f) 34 and the output current node 36 tocompensate for the inductance inherent in the photo diode 32 (e.g.,caused by bond wires within a transistor outline package containing thephoto diode.)

A resistor R1 38 is coupled between a power supply V_(cc) 37 and theinput 30 a of the low gain amplifier 30.

The photo diode's current, I_(pd), is integrated by its own capacitance,C_(pd), to produce a voltage of value V_(pd) where V_(pd) is expressedas: $V_{pd} = {\frac{1}{C_{pd}}{\int{I_{pd}.}}}$

Because this circuit uses positive feedback, the voltage across thephoto diode, V_(pd), is impressed or forced across the feedbackcapacitor C_(f). The voltage across the feedback capacitor, V_(f), istherefore equal to V_(pd) and can be expressed as:$V_{f} = {V_{pd} = {\frac{1}{C_{pd}}{\int{I_{pd}.}}}}$

The voltage V_(f) across the feedback capacitor is differentiated by thefeedback capacitor C_(f) to develop a feedback current I_(f) that can berepresented by the following equations: $\begin{matrix}{I_{f} = {C_{f}\quad \frac{\quad}{t}V_{f}}} \\{= {C_{f}\quad \frac{}{t}\quad \frac{1}{C_{pd}}{\int{I_{pd}{t}}}}} \\{= {\frac{C_{f}}{C_{pd}}{I_{pd}.}}}\end{matrix}$

Therefore, feedback current I_(f) is equal to the photo diode currentI_(pd) amplified by a factor of C_(f)/C_(pd). The current I_(f) producedby the feedback element does not flow back into the photo diode due tothe high input impedance of the low gain amplifier 30. Therefore,current I_(f) flows out of the output current node 36 to a low impedancesecond stage circuit 39 as current I_(out) 36 a. The second stagecircuit 39 balances the voltage accumulation on the feedback capacitorC_(f) 34 due to the photo diode 32 by drawing current from the photodiode amplifier circuit, allowing the voltage across the feedbackcapacitor 34 to be balanced at a continuous and safe level.

The photo diode amplifier circuit of FIG. 3 contributes very littlenoise since the feedback element is a capacitor, which is noiseless.There is no resistor noise since there is no feedback resistor, unlikemost conventional pre-amplifiers. Noise is contributed only by the lowgain amplifier 30. Since the first stage amplifier circuit determinesthe sensitivity of the entire circuit and since noise can not be removedonce it is added, it is especially important for the first stage circuitto add as little noise as possible in order to obtain a highsignal-to-noise ratio.

Even though it is a positive feedback circuit, the photo diode amplifiercircuit will not oscillate if a low gain amplifier 30 with a gain of 1or less is used since the open loop gain of the circuit is less than1.0. As stated above, the open loop gain is the product of the gain ofthe low gain amplifier 30 and the gain of the feedback gain circuit. Thegain of the feedback gain circuit in the photo diode amplifier isexpressed as:${Gain} = \frac{S \times C_{f} \times R}{\left( {S \times C_{f} \times R} \right) + 1}$

where S=2π (frequency), C_(f) is the capacitance value of the feedbackcapacitor, and R is the resistance of the second stage circuit. As theabove equation shows, the gain of the feedback gain circuit will alwaysbe less than one. If a low gain amplifier 30 with a gain of 1 or less isused, such as a unity gain amplifier, the open loop gain at allfrequencies will be less than one and the photo diode amplifier circuit300 will not oscillate.

Referring to FIG. 4, an alternative embodiment of the present inventionis shown wherein a high impedance circuit 400 is coupled to the photodiode amplifier 300. The high impedance circuit 400 has an associatedinput impedance Z_(in), as viewed from input node 40, and is comprisedof a feedback loop that has an input transistor 41 whose gate is coupledto the input node 40 of the low gain amplifier 30 and an outputtransistor Q5 45 whose drain supplies current to the input node of thephoto diode amplifier 300 in accordance with the voltage on the inputnode 40. In a preferred embodiment the high impedance circuit is formedby three p-channel FET transistors 41, 44, 45, two n-channel FETtransistors 42, 43, and a capacitor C 46; however the exact design ofthe high impedance circuit and the number of transistors in it may varyconsiderably in other embodiments.

Referring back to FIG. 3, the high impedance circuit 400 is coupledbetween the power supply 37 and the photo diode amplifier 300 replacingresistor R1 38. The high impedance circuit 400 is used to bias andextend the low frequency cut-off of the photo diode amplifier 300 tolower frequencies (i.e., reduce the low frequency cut-off) than can beachieved using the circuit of FIG. 3. For the photo diode amplifier 300to be biased correctly, the resistance value of resistor R1 38 must below enough to allow adequate voltage from the power supply 37 to reachthe photo diode amplifier 300. However, a low resistance value forresistor R1 38 will not provide an adequate low frequency 3 dB cut-offin the photo diode amplifier, which is determined by the equation:${{low}\quad {frequency}\quad 3\quad {dB}\quad {cut}\text{-}{off}} = \frac{1}{2\quad \pi \quad {R1} \times C_{pd}}$

where C_(pd) is the capacitance value of the photo diode 32. A highresistance value for R1 38 would provide a low cut-off frequency in thephoto diode amplifier 300, but that would be in conflict with therequirement that the resistance value of R1 38 be low enough to allowproper biasing of the photo diode amplifier 300. By replacing resistorR1 38 with the high impedance circuit 400, a high input impedance can beprovided, allowing the low frequency cut-off to be extended to lowerfrequencies while also allowing correct biasing of the photo diodeamplifier 300.

Referring to FIG. 4, a first transistor Q1 41 is coupled to the photodiode amplifier 300 at the gate terminal of transistor Q1 and to a powersupply 47 at the source terminal of transistor Q1. The drain terminal ofthe first transistor Q1 41 is coupled to a capacitor C 46 and to thedrain terminal of a second transistor Q2 42. The second transistor Q2 42is coupled to a third transistor Q3 43 with the source terminals of bothtransistors Q2 42, Q3 43 connected to circuit ground (or other fixedpotential circuit node). The drain of the third transistor Q3 43 iscoupled to the gate and drain terminals of a fourth transistor Q4 44.The fourth transistor Q4 is coupled to a fifth transistor Q5 45, withthe source terminals of both transistors Q4, Q5 connected to the powersupply 47 and the gates of both transistors Q4, Q5 being connected tothe drain terminal of transistor Q3. Transistor Q5 mirrors the currentin transistor Q4, which is governed by transistor Q3 in accordance withthe current produced by the photo diode 32.

The gate terminal of the first transistor Q1 41 senses the voltageproduced at the cathode of the photo diode 32 in the photo diodeamplifier 300 while the fifth transistor Q5 supplies the photo diode 32with current. If the fifth transistor Q5 does not supply the photo diode32 with enough current, the feedback loop formed by transistors Q1 to Q5(41 to 45) forces the fifth transistor Q5 45 to provide more current.

The capacitor C 46 is used to increase the impedance of the highimpedance circuit 400. The impedance Z_(in) looking into the highimpedance circuit 400 is calculated as:${{Z_{in} = \frac{S \times C}{{gm1} \times {gm5}}}}{R5}_{out}$

where S=2π (frequency), gm1 is the transconductance of the firsttransistor Q1 41, gm5 is the transconductance of the fifth transistor Q545, R5 _(out) is the output impedance of the fifth transistor Q5 45, and“A||B” represents the impedance of two parallel impedance elements:$\left. A||B \right. = {\frac{A \times B}{A + B}.}$

Since the variable S varies with frequency, the input impedance Z_(in)of the high impedance circuit 400 also varies with frequency. At lowfrequencies, Z_(in) is approximately equal to (S*C)/(gm1*gm5), while athigh frequencies Z_(in) is approximately equal to R5 _(out). Assumingthe inductive term (S*C)/(gm1*gm5) dominates at low frequencies, the lowfrequency 3 dB cut-off is determined by the equation:${{low}\quad {frequency}\quad 3\quad {dB}\quad {cut}\text{-}{off}} = \frac{1}{2\quad \pi \sqrt{\frac{C \times C_{pd}}{{gm1} \times {gm5}}}}$

where C_(pd) is the capacitance value of the photo diode 32 and C is thecapacitance of capacitor C 46. However, if the capacitance value ofcapacitor C 46 is made large enough, the resistive term R5 _(out) willdominate and the low frequency 3 dB cut-off will be determined by theequation:${{low}\quad {frequency}\quad 3\quad {dB}\quad {cut}\text{-}{off}} = {\frac{1}{2\quad \pi \quad {R5}_{out} \times C_{pd}}.}$

These equations show that the low frequency cut-off of the photo diodeamplifier 300 is governed by the capacitance C of capacitor 46 and theoutput resistance R5 of transistor Q5 of the high impedance circuit 400.Thus, the low frequency cut-off can be explicitly controlled byselecting the size of capacitor 46 and by sizing transistor Q5 tocontrol its output resistance R5.

In an alternative embodiment of the high impedance circuit, one or morecascode transistors can be cascaded in series and coupled between thefifth transistor Q5 45 and the photo diode amplifier 300 to increase theeffective impedance of the fifth transistor Q5 45 and high impedancecircuit 400. The higher input impedance provided by this alternativeembodiment circuit further extends the low frequency cut-off of thephoto diode amplifier 300 to even lower frequencies.

The high impedance circuit 400 allows adequate voltage to reach thephoto diode amplifier 300 from the power supply 47. The voltage beingsupplied to the photo diode amplifier 300 is the voltage at the powersupply 47 minus the gate-source voltage of the sense transistor Q1 41.Since gate-source voltage is normally less than 700 mV in FETtransistors, the voltage supplied to the photo diode amplifier 300 istypically around V_(cc)−700 mV, which is large enough to bias the photodiode amplifier 300 correctly.

Therefore, the high impedance circuit 400 provides high input impedanceZ_(in) at node 40 while still allowing enough voltage to correctly biasthe photo diode amplifier 300. The lower bandwidth extension (i.e., thelow frequency cut-off) of the photo diode amplifier 300 is determined bythe input impedance of transistor Q5 in the high impedance circuit 400,where a larger input impedance will provide a broader bandwidth. Theimpedance looking into the source or drain of a FET is inverselyproportional to the square root of the ratio W/L (where W is the widthof the FET's channel and L is the length of the FET's channel).Therefore, a smaller W/L ratio for transistor Q5 provides a higherimpedance looking into the source or drain of the transistor Q5 (withthe gate at a low impedance).

Lowering the low frequency cut-off of the photo diode amplifier 300enables the photo diode amplifier 300 to amplify slow (i.e., lowfrequency) signals or signals with low frequency content (e.g., signalsthat have long strings of ones or zeroes).

Voltage Source to Current Source Converter

Referring to FIG. 5, a voltage source to current source converter isshown as a further embodiment of the present invention. The voltagesource to current source converter is implemented using a low gainamplifier 50 with an input 50 a and a output 50 b, a floating voltagesource 52, and a resistor R1 54 as a feedback element. The floatingvoltage source 52 and feedback resistor 54 are coupled at an outputcurrent node 56. In an alternative embodiment, the output terminal 50 bof the low gain amplifier 50 is coupled to an output voltage node 58that has an associated output voltage V_(out).

The floating voltage source 52 produces a voltage V_(s) that isimpressed across the resistor R1 54 as voltage V1 due to the feedbacksystem. The voltage across resistor R1 54 produces a current I1 equal toV1/R1. The current I1 does not flow back into the floating voltagesource 52 due to the high impedance of the low gain amplifier 50.Therefore, current I1 flows from the output current node 56 as outputcurrent I_(out) 56 a to a second stage circuit 59, which preferably hasa low input impedance.

The voltage source to current source converter can be used with anysecond stage circuit 59 requiring a current input. At the second stagecircuit 59, the output current I_(out) 56 a can be further amplified toa useable level or it can be used as a current reference or a biascurrent since the output current I_(out) 56 a is a voltage independentcurrent source. For example, in GaAs circuits it is normally difficultto implement a voltage independent current source because the p-typedevices required by such current sources can not be used in circuitsmanufactured using GaAs processes. The voltage source to current sourceconverter of the present invention provides a voltage independentcurrent source that may be implemented using n-type devices and,therefore, may be used in circuits manufactured using GaAs processes.

In an alternative embodiment, the circuit shown in FIG. 5 may beimplemented using a capacitor C coupled between the voltage source tocurrent source converter and the second stage circuit 59 to AC couplethe voltage source to current source converter to the second stagecircuit 59.

In another alternative embodiment, the circuit shown in FIG. 5 may beimplemented using a diode with an associated voltage drop V_(D) coupledbetween the voltage source to current source converter and the secondstage circuit 59 to create a set voltage drop between the voltage sourceto current source converter and the second stage circuit 59.

The circuit shown in FIG. 5 can also be implemented as a voltage sourceamplifier by using output voltage node 58, coupled to the low gainamplifier output 50 b, as the output of the circuit. When implemented assuch, the second stage circuit 59 preferably consists of a resistor R2coupled to circuit ground (or other fixed potential circuit node). Inthis embodiment, a current I_(out) flows through resistor R2 with avalue equal to the current I1 through resistor R1 54 so thatI_(out)=I1=V1/R1. Therefore, the voltage V2 across resistor R2 is equalto (V1/R1)*R2. The voltage V_(out) at the output voltage node 58 isequal to the sum of V1 and V2 and is expressed by the equation:

V _(out) =V 1+(V 1/R 1)*R 2=V 1(1+R 2/R 1).

Since the voltage V_(s) produced by the floating voltage source 52 isequal to V1, the above equation shows that the voltage V_(out) producedat the output voltage node 58 is an amplification of the floatingvoltage source V_(s) by a factor of (1+R2/R1) or ((R1+R2)/R1).

Power Supply Independent Current Source

FIG. 6 shows a power supply independent current source in accordancewith the present invention. The power supply independent current sourcehas a positive feedback system that includes a unity gain amplifier 60(consisting of a single transistor in a preferred embodiment) with aninput or gate terminal 60 a and an output or source terminal 60 b, afirst diode 61 in series with a second diode 62 as a floating “signalsource,” and a resistor R 63 as a feedback element. The second diode 62and the resistor R 63 are coupled at an output current node 64 fromwhich an output current lout 64 a flows to a second stage circuit 69.More generally, the power supply independent current source includes adiode set 61-62 having at least one diode and no more than four diodescoupled in series as the floating source producing a voltage drop V_(D).

The voltage independent current source also includes a power supply node65 (for providing power to the circuit from a power supply) coupled toan input terminal of the unity gain amplifier 60 and to a drain terminalof a first n-channel FET transistor 66. A second n-channel FETtransistor 67 is coupled in series with the first FET transistor 66 andthe diode set 61-62 and has a drain terminal coupled to the inputterminal 60 a of the unity gain amplifier 60. A third n-channel FETtransistor 68 is coupled in series with the diode set 61-62 and has asource terminal coupled to circuit ground (or other fixed potentialcircuit node).

The gate and source terminals of the first FET transistor 66 are shortedto create a current source that provides current to the diode set 61-62and to the second transistor 67. The gate and source terminals of thethird FET transistors 68 are also shorted to create a current sourcethat pulls current away from the diode set 61-62. The gate and drainterminals of the second transistor 67 are shorted to create a voltagedrop to offset the voltage drop of the unity gain amplifier 60.

The voltage drop V_(R) across the resistor R 63 is forced to be equal tothe voltage drop V_(D) across the diode set 61-62 by the feedbacksystem. This can be shown by the following analysis. Referring to FIG.6, the voltage at point W is equal to the sum of the voltage at point Xand the offset voltage of the second FET transistor 67. The voltage atpoint Y is equal to the voltage at point W minus the offset voltage ofthe unity gain amplifier 60. Since the offset voltage of the unity gainamplifier 60 is equal to the offset voltage of the second FET transistor67, the voltage at point Y is equal to the voltage at point X.

The voltage at point Z is equal to the voltage at point X minus thevoltage drop V_(D) across the diode set 61-62. The voltage at point Z isalso equal to the voltage at point Y minus the voltage drop V_(R) acrossthe resistor R 63. Since the voltage at points X and Y are equal, thevoltage drops V_(D) and V_(R) must also be equal.

Therefore, the current I_(R) through the resistor R 63 is equal toV_(R)/R or V_(D)/R. Current signal I_(R) flows into point Z along withthe current signal I_(F1) produced by the first FET transistor 66. Thecurrent signal I_(F3) produced by the third FET transistor 68 flows awayfrom point Z as does the current I_(out) that flows to the second stagecircuit 69. By Kirchhoff's Law, the sum of the current flowing intopoint Z must be equal to the sum of the current flowing out of point Z.Therefore, the sum of the currents I_(R) and I_(F1) must be equal to thesum of the currents I_(out) and I_(F3). Since the first and third FETtransistors 66, 68 are identical, the current signals I_(F1) and I_(F3)produced by the transistors are equal. Therefore, currents I_(R) andI_(out) are equal.

Since current I_(out) is equal to I_(R), current I_(out) is equal toV_(D)/R. Therefore, current I_(out) is independent of the power supplyV_(cc) 65 since it relies only on the voltage drop V_(D) across thediode set 61-62. Generally, the voltage drops supplied by the diodes61-62 vary much less (in response to changes in temperature, supplyvoltage, manufacturing process variations, etc.) than voltage dropssupplied by transistors. Since current I_(out) relies only on diodevoltage drops, it is also free from large variations.

The power supply independent current source of the present invention isespecially useful in circuits manufactured using GaAs processes. Asexplained above, a power supply independent current source is normallydifficult to build since the p-type devices normally required by suchcurrent sources cannot be used in GaAs circuits. The power supplyindependent current source of the present invention uses only n-typedevices and therefore may be made entirely from GaAs circuit components.

The power supply independent current source of the present invention canbe used with any second stage circuit 69 requiring a current input. Atthe second stage circuit 69, the output current I_(out) 64 a can befurther amplified or it can be used as a current reference or a biascurrent since the output current I_(out) 64 a is a power supplyindependent current source.

Alternative embodiments of the present invention can also be created byusing one diode or three or more diodes in place of the two diodes inthe diode set 61-62 for the floating signal source. Alternativeembodiments can also be created by using other types of low gainamplifiers in place of the unity gain amplifier 60.

Embedded/Nested Positive Feedback Circuit Stages

The signal processing circuits of the present invention can also beembedded or nested in multiple stages of signal processing to acquiremultiple stages of gain or to create more complex functions. FIG. 7shows an example of how the voltage source amplifier of FIG. 5 can beembedded to produce another stage of gain. The embedded voltage sourceamplifier includes an inner voltage source amplifier 80 and an outervoltage source amplifier 82.

The inner voltage source amplifier 80 includes a first low gainamplifier 70 with an input 70 a and an output 70 b, a floating voltagesource 71 producing a voltage V_(s), a first resistor R1 73 as afeedback element, and a second resistor R2 74 as an inner second stagecircuit. The floating voltage source 71 and first resistor 73 arecoupled at a first output current node 72 that is coupled to the secondresistor R2 74. The output 70 b of the first low gain amplifier 70 iscoupled to a first output voltage node 77 that has an associated outputvoltage V_(out1).

The outer voltage source amplifier 82 includes a second low gainamplifier 75 with an input 75 a and an output 75 b, the inner voltagesource amplifier 80 as a floating voltage source producing a voltageV_(out1), a third resistor R3 78 as a feedback element, and a fourthresistor R4 79 as a outer second stage circuit. The inner voltage sourceamplifier 80 and the third resistor R3 78 are coupled at a second outputcurrent node 77 (which is also the first output voltage node 77). Thefourth resistor R4 79 is coupled to the second output current node 77and coupled to circuit ground (or other fixed potential circuit node).Alternately, the fourth resistor R4 is coupled to an additional embeddedcircuit instead of being coupled to the circuit ground. The outervoltage source amplifier 82 provides the additional embedded circuit acurrent signal that is an amplified version of the input signal from thefloating voltage source 71. The output 75 b of the second low gainamplifier 75 is coupled to a second output voltage node 76 that has anassociated output voltage V_(out2).

As shown above, each voltage source amplifier amplifies the floatingvoltage source signal by a gain factor of ((R_(x)+R_(y))/R_(x)) whereR_(x) is the feedback resistor and R_(y) is the second stage circuitresistor that is coupled to the floating voltage source and the feedbackresistor R_(x). Therefore, the inner voltage source amplifier produces again factor of ((R1+R2)/R1) and the outer voltage source amplifierproduces a gain factor of ((R3+R4)/R3). The total gain of the embeddedcircuit as measured by the ratio V_(out2)/V_(s) would be the product ofthe two gain factors or ((R1+R2)/R1)*((R3+4)/R3).

Several embodiments of the present invention may be implemented usinggallium arsenide (GaAs) metal-semiconductor field-effect transistors(MESFETs). In other embodiments, the circuits and signal processingmethods of the present invention may be implemented using bipolar andcomplementary metal-oxide silicon or BiCMOS circuitry.

Alternative embodiments of the present invention can also be created byembedding three or more signal processing circuits to obtain greatergain factors or more complex functions.

Alternative Embodiments Current Source to Current Source Converter

FIG. 8 shows a diagram of an alternative embodiment of the floatingsignal source 22 shown in FIG. 2. An alternative source 22′ has acurrent source 84 producing a current I_(s) and a resistor R2 86. Thecurrent source 84 is coupled to the resistor R2 86 on one end andcoupled to circuit ground (or other fixed potential circuit node) on itsother end. The alternative source 22′ is used in the positive feedbackcircuit of FIG. 2 to implement a current source to current sourceconverter. The current source to current source converter also includesa resistor R1 as a feedback element (24 of FIG. 2). The current sourceto current source converter produces an output current I_(out) (26 a ofFIG. 2) that is equal to the current source current I_(s) amplified by afactor of R2/R1 and can be represented by the equation:I_(out)=(R2/R1)*I_(s).

Integrator Circuit

FIG. 9 shows a diagram of an alternative embodiment of the floatingsignal source 22 shown in FIG. 2. An alternative source 22″ has avoltage source 96 producing a voltage V_(s) and a capacitor C 94. Thecapacitor C 94 is coupled to the voltage source 96 on one end andcoupled to circuit ground (or other fixed potential circuit node) on itsother end. The alternative source 22″ is used in the positive feedbackcircuit of FIG. 2 to implement an integrator circuit. The integratorcircuit also includes a resistor R as a feedback element (24 of FIG. 2).The integrator circuit produces an output voltage V_(out) at the outputvoltage node (27 of FIG. 2) that is equal to the integration of thevoltage source voltage V_(s) divided by (R*C), where R is the resistanceof the feedback element 24 and C is the capacitance of capacitor 94, andcan be represented by the equation:$V_{out} = {\frac{1}{RC}{\int{{Vs}.}}}$

Differentiator Circuit

FIG. 10 shows a diagram of an alternative embodiment of the floatingsignal source 22 shown in FIG. 2. An alternative source 22′″ has avoltage source 106 producing a voltage V_(s) and a resistor R 104. Theresistor R 104 is coupled to the voltage source 106 on one end andcoupled to circuit ground (or other fixed potential circuit node) on itsother end. The alternative source 22′″ is used in the positive feedbackcircuit of FIG. 2 to implement a differentiator circuit. Thedifferentiator circuit also includes a capacitor as a feedback element(24 of FIG. 2). The differentiator circuit produces an output voltageV_(out) at the output voltage node (27 of FIG. 2) that is equal to thedifferentiation of the voltage source voltage V_(s) multiplied by (R*C),where R is the resistance of resistor 104 and C is the capacitance ofthe capacitor feedback element 24, and can be represented by theequation: $V_{out} = {{RC}\frac{}{t}{{Vs}.}}$

What is claimed is:
 1. A signal processing circuit, comprising: an innerstage circuit comprising: a first low gain amplifier having an input andan output; and a first positive feedback circuit coupling the output ofthe first low gain amplifier to the input of the first low gainamplifier; the positive feedback circuit including a floating signalsource, a first feedback element coupled in series with the floatingsignal source at a first output current node, and an inner second stagecircuit coupled to the first output current node; the floating signalsource producing a first voltage that is impressed across the firstfeedback element and processed by the first feedback element into afirst output current signal; and an outer stage circuit coupled to theinner second stage circuit of the inner stage circuit for receiving thefirst output current signal therefrom, the outer stage circuitcomprising: a second low gain amplifier having an input and an output;and a second positive feedback circuit coupling the output of the secondlow gain amplifier to the input of the second low gain amplifier; thesecond positive feedback circuit including the inner stage circuit, asecond feedback element coupled in series with the inner stage circuitat a second output current node, and an outer second stage circuitcoupled to the second output current node; the inner stage circuitproducing a second voltage that is impressed across the second feedbackelement and processed by the second feedback element into a secondoutput current signal.
 2. The signal processing circuit of claim 1,wherein the first feedback element is comprised of a resistive elementand the second feedback element is comprised of a resistive element. 3.The signal processing circuit of claim 1, wherein the first low gainamplifier has a gain of at least 0.5 and no more than
 2. 4. The signalprocessing circuit of claim 1, wherein the second low gain amplifier hasa gain of at least 0.5 and no more than
 2. 5. The signal processingcircuit of claim 1, wherein the inner second stage circuit is comprisedof a resistive element and the outer second stage circuit is comprisedof a resistive element.
 6. The signal processing circuit of claim 1,wherein the floating signal source is a photo diode that produces acurrent I_(pd) and has an associated capacitance, C_(pd).
 7. The signalprocessing circuit of claim 1 including: a resistor coupled between thesecond output current node and the outer second stage circuit.
 8. Thesignal processing circuit of claim 1 including: a capacitor coupledbetween the second output current node and the outer second stagecircuit.
 9. The signal processing circuit of claim 1 including: a diodecoupled between the second output current node and the outer secondstage circuit.